1. Field of the Invention
The present invention relates to a CDMA (Code Division Multiple Access) communication system and more particularly to a reverse spreading device being capable of performing timing detection and channel estimation even in a large frequency offset environment.
2. Description of the Related Art
Recently, a CDMA communication system being highly tolerant of interference is widely used as on of the communication methods employed in a mobile communication system. The CDMA communication system is a communication method in which a sender transmits a user signal after spreading it using a spread code (one-bit signal) and a receiver obtains an original user signal by reversely spreading a received signal using a same spread code as used on a sender side. Because of his, the received signal cannot be reversely spread by the receiver unless a phase of the spread code string of the receiver is synchronized to that of the spread code string used by the sender. To achieve this, at a mobile station, TCXO (Temperature Compensated Crystal Oscillator) having a high frequency precision is used as a reference oscillator used to generate a reference frequency signal required for demodulation of a signal received from a base station. However, since the mobile station must be made small and be highly cost-effective, it is natural that a frequency precision of a reference oscillator used for the mobile station is lower than that of the reference oscillator used for the base station.
Therefore, an AFC (Automatic Frequency Control) is exercised at the mobile station in order to match frequency of the reference frequency signal to that of the reference frequency signal sent from the base station by the sender.
Configurations of the mobile station in which the AFC is exercised will be described by referring to FIG. 12. In the following description, it is assumed that, in the CDMA communication system, one symbol signal is spread by using spread codes of n-pieces of chips.
The mobile station in FIG. 12 is composed of a radio section 1, a timing detecting device 4, a channel estimating device 5, a TCXO 6, a demodulating section 16 and an AFC circuit 17. The timing detecting device 4 has a reverse spreading device 7 and a peak detecting section 8. The channel estimating device 5 has a reverse spreading device 9 and a rotation correcting section 15.
The radio section 1 is adapted to perform quadrature detection of a received high frequency signal based on a reference frequency signal generated by the TCXO 6 and carries out an analog-digital conversion so that the received high frequency signal is converted to a base band signal 11 composed of a digital signal I component (meaning In-phase signal) and a base band signal 21 composed of another digital signal Q component (meaning Quadrature-phase signal).
TCXO 6 is operated to output the reference frequency signal whose frequency is controlled by the AFC circuit 17, as the reference frequency signal. The reverse spreading device 7 performs reverse spreading by multiplying the base band signal 11 and base band signal 21 one being composed of the I component and another being composed of the Q component fed from the radio section 1, by the spread code.
The peak detecting section 8 detects spreading timing which is timing in which a correlation value reaches a peak level when the reverse spreading is performed by the reverse spreading device 7. The reverse spreading device 9 is adapted to obtain a complex symbol by performing the reverse spreading of the base band signal 11 and base band signal 21 composed respectively of the I component and Q component fed from the radio section 1 using the spreading timing obtained by the peak detecting section 8.
Configurations of the reverse spreading devices 7 and reverse spreading device 9 implemented by using digital matched filters will be described by referring to FIG. 13.
Each of the reverse spreading device 7 and reverse spreading device 9 is composed of a correlator 110 and a correlator 120 respectively. The correlator 110, if an over-sampling is made at a time of receiving a signal, is composed of OSR x (n−1) pieces of delay devices 21 to 12OSR (n−1), where the “OSR” represents an “over-sampling ratio” employed at the time of receiving the signal, n-pieces of multipliers 131 to 13n and an adder 14. Similarly, the correlator 120 is composed of OSR x (n−1) pieces of delay devices 22, to 22OSR(n−1), n-pieces of multipliers 231 to 23n and an adder 24.
The correlator 110 is adapted to calculate a correlation value by allowing an inputted base band signal 11 composed of the I component to be shifted sequentially through the delay devices 121 to 12OSR(n−1) and to be multiplied sequentially by the spread code. The adder 14 calculates the correlation value containing the I component by adding each individual correlation value obtained sequentially to another individual correlation value. Similarly, in the correlator 120, a correlation value composed of the Q component is obtained by reversely spreading a base band signal 21 composed of the Q component using the spread code. The pair of correlation values containing the I component and Q component becomes a reversely spread complex symbol.
Configurations of the reverse spreading device 7 and reverse spreading device 9 implemented by using sliding correlators will be described by referring to FIG. 14.
Each of the reverse spreading device 7 and reverse spreading device 9 is composed of a correlator 60 and a correlator 70. The correlator 60 has a multiplier 62, an adder 63 and a delay device 64. The correlator 70 has a multiplier 72, an adder 73 and a delay device 74.
The correlator 60 is adapted to multiply an inputted base band signal 11 composed of the I component by a spread code for every chip using the multiplier 62. The n-pieces of values obtained by multiplying the resulting base band signal 11 by the spread code are integrated by an integrator composed of the adder 63 and delay device 64 to produce a correlation value containing the I component. Similarly, the correlator 70 is adapted to calculate a correlation value containing the Q component by reversely spreading the base band signal 21 composed of the Q component. The pair of correlation values containing the I component and Q component become the reversely spread complex symbol signal.
The rotation correcting section 15 (FIG. 12) detects a phase error contained in the complex symbol of the I component and Q component obtained from the reverse spreading device 9 and corrects the phase error.
Next, a channel estimation performed by the rotation correcting section 15 in the channel estimating device 5 will be described below.
Channel estimation represents processes of estimating a phase of the complex symbol which has been rotated due to frequency offset of the reference frequency signal at the mobile station or a like and of correcting the phase. The channel estimation is carried out by using a pilot symbol contained in data sent from the base station as a reference.
First, the pilot symbol is described. Frame configurations, designated according to a specification, of a perch channel of a forward link through which signal is sent from the base station to the mobile station are described below by referring to FIG. 15.
A 720 ms super frame constituting the perch channel is composed of 72 pieces of 10 ms radio frames 501 to 5072. Each of the radio frames 501 to 5072 contains 16 pieces of time slots 511 to 5116. Each of the time slots 501 to 5016 includes a search code symbol 52 composed of one symbol, a sending data symbol 53 composed of five symbols and a pilot symbol 54 composed of four symbols. Though the pilot symbol 54 has a different value for each of the time slots 501 to 5016, its pattern is a predetermined pattern. Therefore, the mobile station can get information about the pattern of the pilot symbol 54 before the mobile station receives the pilot symbol 54. In the case of the perch channel frame configurations described above, the mobile station can make measurement of phase error and frequency error in signals sent by the base station by using the pilot symbols 54 with four symbols.
Four complex symbols constituting the pilot symbol 54 are plotted on a plane with the Q component as an ordinate and with the I component as an abscissa. If complex vectors 45, 46, 47 and 48 are given as shown in FIG. 16, there is a phase rotation by θ1 between the complex vector 45 and complex vector 46, by θ2 between the complex vector 46 and complex vector 47 and by θ3 between the complex vector 47 and complex vector 48.
The demodulating section 16 (FIG. 12) is adapted to obtain an original symbol by demodulating the complex symbol composed of the I and Q components, the phase error of which is corrected by the rotation correcting section 15. The AFC circuit 17 is operated to calculate frequency error which is a difference between frequency of the reference frequency signal generated by the TCXO 6 and reference frequency of signal from the base station and to control the frequency of the reference frequency signal generated by the TCXO 6 so as to reduce frequency errors.
At the conventional mobile station in which an AFC method is performed, even when the phase of the complex symbol obtained by the reverse spreading device 9 is rotated due to occurrence of frequency offset in the reference frequency signal generated by the TCXO 6, if the frequency offset of the reference frequency signal is with in a range, the phase error is corrected by the rotation correcting section 15 and normal modulation is made by the demodulating section 16.
However, at the conventional mobile station described above, if the frequency offset of the reference frequency signal exceeds the predetermined range, the frequency offset cannot be corrected by the AFC method. That is, the frequency offset may exceed the range being within the AFC.
Reasons why such phenomena as above occur in the conventional mobile station will be described below.
If carrier frequency offset occurs, phase is rotated in n-chip area being one symbol area. That is, phase error occurs among chips.
However, in the conventional reverse spreading device 7, the complex symbol is obtained by calculating the correlation value in a state in which all signals of n-pieces of chips in one symbol area are in phase. Because of this, the correlation value obtained by adding the correlation value of each chip to that of the spread code is made small, thus causing a decrease in spread gain. This also causes a probability of correct synchronization being a probability of getting spreading timing to be decreased in the timing detecting device 4. If the spreading timing cannot be obtained, not only reverse spreading by the reverse spreading device but also subsequent channel estimation and AFC process are made impossible as well.
Next, a state in which the probability of correct synchronization decreases with increase in the frequency offset is explained by referring to FIG. 17. FIG. 17 is a graph showing a relationship between the probability of correct synchronization and energy versus mean noise power spectrum density (Eb/N0) per one bit of a signal when an amount of a frequency offset of a carrier frequency is used in the conventional reverse spreading device.
It is apparent from the graph that, when the frequency offset is 0 (zero) ppm, the probability of correct synchronization does not decrease even if the Eb/N0 is decreased, while the probability of correct synchronization decreases when the frequency offset increases to become 3 ppm and 5 ppm. When the frequency offset becomes 5 ppm in particular, the probability of correct synchronization decreases rapidly.
Moreover, in the channel estimation by the channel estimating device 5 (FIG. 12), if spreading gain decreases, an error rate increases. In the conventional mobile station, since the channel estimation is performed based on symbol rate, if phase offset by over 180° per one symbol occurs, the spreading gain decreases greatly and the channel estimation in units of the symbol becomes very difficult. Furthermore, since detection in which direction the phase offset has occurred is impossible, the estimation of the frequency offset amount is also impossible. For example, when a reference frequency of the TCXO 6 is 2 GHz and the symbol rate is 16 Ksps (symbol/second), the offset of the reference frequency by 1 ppm causes phase error of 45°/symbol and offset of the reference frequency by 4 ppm causes the phase error of 180°/symbol.
FIG. 18 is a graph showing a relation between BER (Bit Error Rate) and Eb/N0 per one bit of a signal obtained when an amount of a frequency offset of a carrier frequency is used as a parameter in the conventional spreading device. As apparent from this graph, the BER obtained by the same Eb/N0 increases as the frequency offset increases from 0 ppm to 4 ppm.
Even in the case of the conventional examples described above, if the frequency offset of the TCXO 6 is within a range of ±4 ppm, since amount of the frequency offset can be detected, the offset of the frequency can be corrected by the AFC. This shows that actual range being within the AFC for the frequency offset is about ±4 ppm.
To solve the above problem, technology in which the range of the frequency offset being within the AFC for the frequency offset is expanded is disclosed in Japanese Patent Application Laid-open No. Hei9-200081. FIG. 19 is a schematic block diagram showing configurations of a frequency error detecting circuit containing other conventional reverse spreading devices.
The conventional frequency error detecting circuit is composed of complex matched filter 131 and complex matched filter 132, complex spread code generating device 133 and complex spread code generating device 134, peak detection averaging section 135 and peak detection averaging section 136, a peak position detecting section 137, power calculating section 138 and power calculating section 139, a normalizing circuit 141, a power difference calculating section 143 and a frequency error converting section 142. The power difference calculating section 143 includes an adder 140 and the normalizing circuit 141.
In the conventional frequency error detecting circuit, a base band complex signal is received by a radio section and undergoes quadrature detection and is input into the complex matched filter 131 and then is multiplied by a complex code generated by the complex spread code generating device 133 to produce a complex correlation value and another base band complex signal is also received by the radio section and undergoes quadrature detection and is input into the complex matched filter 132 at a same time and then is multiplied by a complex spread code generated by the complex spread code generating device 134 to produce the complex correlation value. Each of the produced correlation values is averaged by each of two peak detection averaging sections, peak detection averaging section 135 and peak detection averaging section 136, respectively with maximum timing of the complex correlation detected by the peak position detecting section 137 during several symbol times to calculate a power value by the two power calculating sections, power calculating section 138 and power calculating section 139. In the power difference calculating section 143, a difference of the calculated power values is computed by the adder 140 and the resulting difference of power values is normalized by the normalizing circuit 141. The frequency error converting section 142 calculates a corresponding frequency error from difference of a normalized power value and outputs it. The complex spread code generating device 133 outputs a complex spread code calculated in advance by being given a positive frequency offset and the complex spread code generating device 134 outputs a complex spread code calculated in advance by being given a negative frequency offset having a same absolute value as the positive frequency offset. This allows the frequency offset to be given to a complex correlation value within a symbol signal area.
In the conventional frequency error detecting circuit 6, by performing the reverse spreading using the complex spread code obtained by being given the frequency offset in advance, the range being within the AFC for the frequency error can be expanded. However, to achieve this, a memory used to store the complex spread code obtained by being given the frequency offset in advance is required. Moreover, a high speed chip rate processing is also required in which the base band signal can be multiplied by the spread code. In the conventional reverse spreading device as shown in FIG. 13 and FIG. 14, since the spread code is composed of one-bit signal, multiplication of the base band signal by the spread code is carried out actually by code operations.
However, since expression of the spread code obtained by being given the frequency offset requires increased number of bits and since the multiplication of the base band signal by the spread code cannot be achieved by such code operations, a multiplier is required by which signals composed of a plurality of bits can be multiplied by each other. Therefore, circuit scale and power consumption of the reverse spreading device used to do complex multiplication become much larger when compared with a case in which base band signal is multiplied by spread code containing no frequency offset. Such increased circuit scale and power consumption of the reverse spreading device are contradictory to recently increasing requirements for miniaturization and low power consumption in the mobile station such as a portable telephone or a like receiver/transceiver.
The conventional reverse spreading device and the AFC method described above have the following problems:
(1) The timing detection and channel estimation in large frequency offset environments cannot be performed only by the AFC method using the phase error obtained by channel estimation at the symbol rate, resulting in a narrow range being within the AFC for frequency error.
(2) Greatly increased circuit scale and power consumption of the reverse spreading device are inevitable in the AFC method disclosed in the Japanese Patent Application Laid-open No. Hei9-200081.